The KD2BD 9600 Baud Modem

By: John A. Magliacane, KD2BD

This paper was originally published in the February, March, and April 1998 issues of Satellite Times magazine.


The KD2BD 9600 Baud Modem is a low-cost, high-performance 9600 bit per second FSK (Frequency Shift Keying) modem designed to interface between a standard packet radio terminal node controller (TNC) and an FM voice transmitter and receiver. The modem uses commonly available components, allows full-duplex access to the 9600 baud digital communication satellites, and is suitable not only for digital satellite communications, but for terrestrial packet radio communications as well.

The KD2BD 9600 Baud Modem went from conception to reality in just seven days. It uses some of the time-proven signal processing techniques used in the 1200 Baud KD2BD Pacsat Modem developed several years ago, and should be of interest to amateurs wishing to add 9600 baud digital communication capabilities to their satellite or terrestrial packet radio stations.

Design Goals

The KD2BD 9600 Baud Pacsat Modem was designed with several important design goals in mind. First, it was designed to use commonly available components and not rely on special EPROMS for transmit waveform synthesis or bit clock detection, thereby allowing easy, inexpensive, and uncomplicated duplication. Secondly, since it has been shown that even randomized 9600 baud baseband data contains a DC component that is ignored in many other 9600 baud modem designs, DC coupling is used throughout the modulator and demodulator sections of the KD2BD 9600 Baud Modem for optimum performance. And lastly, the design is essentially uncomplicated, allowing an understanding and appreciation of its operation by both veteran OSCAR users and beginners alike.

9600 Baud Communications

Before getting into the details of this particular modem design, it is helpful to look back to the beginning of amateur packet radio communications to have an understanding of the events and logic that led to the protocol standards used today in 9600 baud digital communications. When AX.25 protocol amateur packet radio communications first began in the early 1980s, early experimenters used Bell-202 type audio frequency shift keying (AFSK) telephone modems to pass packet binary data over the air using voice-grade VHF narrowband FM transceivers. Bell-202 modems were selected not for technical or performance reasons, but because of their wide availability at the time on the surplus market. A data rate of 1200 bits per second was used, which was four times faster than what telephone modems were capable of at the time, and was many times faster than the fastest amateur radio radioteletype (RTTY) communications. Although Bell-202 modems were originally designed for 300 bit per second data communications over telephone lines, they were found to function satisfactorily for half-duplex radio work at four times their design speed.

When terminal node controllers made their appearance on the commercial market, they included internal Bell-202 AFSK modems for communications at 1200 bit per second as standard equipment. This was done for compatibility with the early standards set by the packet experimenters, and because the Bell-202 AFSK modem protocol was proven to work satisfactorily for packet communications. Unfortunately, the trend caught on, and while the transmission rate of telephone modems soared from 300 bits per second to 56 kilobits per second, packet radio communications stagnated at just 1200 bits per second, causing many hams to turn away from wireless digital communications in favor of landline-based (Internet) communications.

While some vendors offered packet radio TNCs with 2400 bit per second capabilities, few people purchased them for fear of being incompatible with the rest of the packet community. Before 9600 bps digital satellites came into play, the only people who got involved with higher data rates were those enterprising individuals who developed high speed packet radio "backbone" networks on UHF frequencies. Unfortunately, few end users got the chance to experience high-speed packet radio communications first hand, and directly witness the significant improvement that could be made over a "standard" TNC with an internal Bell-202 modem.

The Bell-202 AFSK modem standard that is still widely used for the majority of 1200 bit per second VHF-FM packet radio communications uses audio tones of 1200 Hz and 2200 Hz to represent the binary '1's and '0's of packet radio's HDLC baseband serial data. The use of an audio frequency shift keying protocol allows binary data to pass over an AC-coupled voice grade communications link. The method works, but its not without its problems. It occupies a lot of bandwidth -- so much so that it is possible to pass data at 9600 bits per second in slightly less bandwidth if a different modem and RF Frequency Shift Keying (FSK) modulation are used.

At 9600 bits per second, it is not possible to convert the '1's and '0's of a serial binary data stream into audio tones for application to an FM voice transmitter and remain within legal RF bandwidth limitations. The use of AFSK at 9600 bits per second would also exceed the bandwidth limitations of voice-grade communications equipment. Instead, 9600 bit per second baseband data is used to directly frequency modulate the RF carrier of the transmitter.

When early high-speed packet radio communications experiments were conducted on VHF frequencies over a decade ago, Steve Goode, K9NG, found that directly modulating an FM transmitter with transmit data generated by a packet radio terminal node controller was not a very effective way of transmitting high-speed data. To begin with, the square waveshape of the TTL-level data generated by the TNC is rich in harmonics and would occupy a very wide bandwidth if used to directly modulate an RF carrier. Steve found that passing the transmit data stream through a multi-section low-pass filter was an effective way of reducing the bandwidth of the transmitted signal to the minimum required for effective communications. The smaller bandwidth also reduces the chances of causing interference to adjacent channel users, and allows the signal to pass through the IF filters of standard narrowband FM voice receivers without excessive phase distortion.

Steve also realized that the unsymmetrical waveshape of AX.25 baseband data carried a significant DC component that needed to pass without distortion through the RF communications link between transmitter and receiver. In theory, this is not difficult. In reality, however, the frequency stability of commonly available narrowband VHF-FM communications equipment was found to be rarely high enough to pass a DC-referenced digital signal without significant bias distortion. Something needed to be done to reduce the DC component level of the signal in such a way that would not increase its bandwidth.

Steve tackled this problem by passing all transmitted data through a scrambling or randomization circuit prior to transmission and unscrambling received data to restore the signal back to normal. The scrambler randomized the data transmitted data pattern, thereby minimizing the chances of transmitting long runs of '1's or '0's or repetitious patterns containing significant low frequency energy.

Randomizing techniques are typically identified by the scrambling polynomial they synthesize. Steve's circuit was based on a 17-bit maximal length linear feedback shift register (LFSR). A total of eight different maximal length randomizing techniques can be employed using a single tap 17-bit shift register. Steve's design used a scrambling polynomial of 1+x^12+x^17, whereas old 1200 baud Bell-212A telephone modems used a scrambling polynomial of 1+x^14+x^17. The technique Steve chose has become the standard for 9600 bit per second digital communications, and is authorized by the Federal Communications Commission for amateur use. The technique produces a pseudo-random data sequence of bits that repeats after 131,071 clock pulses, or every 13.65 seconds at 9600 bits per second. A maximum of 17 ones or 16 zeros can occur in a row with this method of scrambling. Figure 1 illustrates the design of the type of data randomizer used in amateur communications.

Circuit Overview

Figure 2 shows a block diagram of the KD2BD 9600 Baud Pacsat Modem. The modem consists of a data modulator and demodulator that act independently of one another. This mutual exclusion allows the modem to be used for full-duplex satellite communications as well as half-duplex (simplex) terrestrial communications.

The modulator portion of the modem is very simple. It takes transmit data (TXD) and clock signals (TXCLK) from the host TNC, applies them to a 17-bit linear feedback shift register scrambler, and filters the resultant through a 6th order low-pass filter. This technique produces an exceptionally clean raised cosine waveshape that is suitable for direct connection to the modulator of an FM voice transmitter.

The demodulator portion of the modem is a bit more complicated. The demodulator connects to the detector of a narrowband FM voice receiver, and gently filters the received signal through a second order low-pass filter. The purpose of the filter is remove any residual 455 kHz IF noise that may be present on the demodulated signal. The filtered signal is then fed to a pair of precision peak voltage detectors that are used to measure the maximum positive and negative voltage excursions of the input waveform. The average of the peak excursions represents the voltage midpoint of the received waveshape regardless of any DC offset present on the received signal, and is used as a reference in a data slicer and a digital automatic frequency control (AFC) circuit. The data slicer is used to convert the received signal to a square waveshape for later processing in digital logic circuits. The digital AFC is used to slowly tune the receiver lower in frequency in compensation for Doppler shift when receiving signals from satellites in low earth orbit.

The processed signal is then diverted into two different directions. The first direction takes it through a bit clock regenerator, and the second takes it through an unscrambler and then out to the host TNC for packet disassembly.

Bit clock regeneration is performed in a very uncomplicated manner. The filtered and processed received signal is applied to an edge detector designed around an exclusive-OR (XOR) gate and an RC delay network. The edge detector produces a short output pulse each time the waveform of the received signal passes through its voltage midpoint. The waveform produced by the edge detector is known as a protoclock, and is in phase with the clock of the received signal. Unfortunately, the waveform produced is non-continuous, so it is processed through a phase locked loop (PLL) circuit. The phase locked loop operates at the bit rate of the incoming signal (9600 Hz), and rapidly locks its oscillator to the average phase of the protoclock, thereby generating a continuous, noise-free clock suitable for further processing of the received signal.

The received signal is also passed through a fifth-order low-pass filter. The filter serves two purposes. First, it attenuates any remaining noise present on the received signal, and secondly, it delays the incoming signal long enough so that the center of each received bit is concurrent with the rising edge of the regenerated clock waveform. This delay is necessary because the unscrambler is triggered on the rising edge of the regenerated clock signal rather than the center of each clock pulse. The filtered signal is then applied along with the regenerated clock to the unscrambler, the output of which is converted to TTL levels and made available to the host TNC for final processing.

Filter Design

The low-pass filters used to remove noise from the received signal and properly shape the waveform of the transmitted signal play a critical role in the overall performance of the KD2BD 9600 Baud Modem. Considerable time and effort was spent in testing and optimizing these filters to adequately process the received and transmitted waveforms without introducing undesirable products, such as phase distortion. The filters need to have a flat frequency response up to their cutoff frequency, and must also exhibit a flat group delay characteristic. Uneven frequency response causes an unfair bias either toward or away certain bit pattern sequences, while uneven group delay causes the filtered waveform to undergo uneven propagation time depending on the frequency of the signal being passed through the filter. In terms of data communications, uneven group delay produces phases jitter which causes bit zero crossing point instability, making receiver bit clock detection and extraction difficult and less reliable.

Figure 3 shows the frequency response and group delay characteristics of the receive filter used in the G3RUH 9600 Baud Modem (Issue 3) as simulated using MicroCAP IV circuit analysis software. Notice the uneven low frequency response and the excessive group delay distortion below 400 Hz. While the purpose of the receive filter is to gently remove IF noise from the received signal, the analysis suggests that the G3RUH receive filter may actually do more harm than good to the incoming signal. On the transmit side, the G3RUH Modem uses a Transversal or Finite Impulse Response (FIR) filter to tailor the transmit waveshape to compensate for distortions present in the IF filters typically used in commonly available FM receivers.

Figures 4 and 5 show the simulated frequency response and group delay characteristics of the receive and transmit filters used in the TAPR/K9NG 9600 Baud Modem designed in August 1985. The TAPR/K9NG filters offer a much flatter frequency response, but the receive filter is still a bit narrow, and suffers from some low frequency roll-off and group delay distortion.

Figures 6 and 7 show the simulated frequency response and group delay characteristics of the receive and transmit filters used in the KD2BD 9600 Baud Modem. The extremely flat frequency response and constant group delay characteristics of these filters produce an exceptionally clean raised-cosine transmit waveform and introduce virtually no distortion to the received signal. These filters were actually designed by empirically selecting component values that produced the cleanest waveform patterns as viewed on an oscilloscope. It was only after the filters were designed and the modem constructed that the computer simulations were run, effectively confirming their performance and characteristics.

Receiving 9600 Baud FSK

9600 baud Frequency Shift Keying (FSK) modems are typically used in conjunction with communications quality FM receivers and transmitters. In actuality, 9600 baud modems are signal processors more than they are modems (Modulator/Demodulators) since the actual signal modulation and demodulation processes occur in the groundstation radio equipment rather than in the electronics that is typically referred to as a "modem".

One of the radical features of the KD2BD 9600 Baud Modem design is that it uses full DC coupling to the receiver's detector and the transmitter's modulator. This direct coupling provides a frequency response that extends for the full spectrum of the FSK signal. DC coupling to an FM receiver's discriminator and transmitter's varactor diode requires some careful design considerations. First, let's examine how 9600 baud FSK signals are received and demodulated.

FM Detectors

Demodulators used in receivers for FM detection exhibit an "S Shaped" response curve. Figure 8 shows the response of a typical FM demodulator. The demodulator's linear curve extends for several kilohertz above and below the center of the receiver's final IF frequency. As can be seen in the figure, an unmodulated carrier applied to an FM demodulator at its center frequency will produce an output voltage of exactly zero volts. A carrier either above or below the center frequency will produce either a positive or negative output voltage. Modulation applied to the frequency of the carrier produces an AC output voltage from the demodulator.

Provided the modulation waveform is symmetrical and the carrier frequency is properly centered, the AC output voltage from the discriminator is centered about the zero volt level. If the received carrier is tuned to one side of the response curve rather than the center, the audio output from the detector will not be centered about the zero level, but instead be centered about a small positive or negative voltage. This voltage is known as an offset voltage, and in some receivers is used for automatic frequency control (AFC) purposes or to drive a zero-center tuning meter.

DC Offsets and Level Correction

Modern FM communications receivers typically use a quadrature detector chip for FM signal demodulation. Quadrature detectors posses the same "S Shaped" response curve as do classic discriminators, but because they are designed using active components powered by a single-ended power supply, quadrature detector output voltages are typically centered about a voltage level that is several volts above ground. A positive DC offset voltage is always present, even with properly tuned signals, and this voltage and varies linearly with receiver tuning.

Even some older transceivers such as the Yaesu FT-726R that use a ceramic discriminator for FM detection elevate the reference of the discriminator several volts above ground so that the output voltage is always positive. With this in mind, we are pretty well guaranteed that the detector output of an FM receiver tuned to an FSK data transmission will consist of an FSK waveform riding on top of a positive DC offset voltage (FSK+DC).

Since the position of the FSK carrier within the passband of the FM detector will vary the offset voltage, the offset will continuously vary over time when receiving transmissions from communication satellites due to the effects of doppler shift. Therefore, if a frequency response down to DC is required and direct coupling is used between the FM receiver detector and the data demodulator portion of an FSK modem, adequate measures must be taken to properly handle the DC offset voltage present on the output of the detector so that it does not interfere with post detection signal processing of the received FSK data.

The first schematic shows the circuitry used to initially process FSK signals received by a narrowband FM receiver. Received signals consisting of a demodulated FSK signal plus a DC offset voltage (FSK+DC) are first processed through a second order low-pass filter designed around operational amplifier UlA. The purpose of the filter is to attenuate any intermediate frequency (IF) noise that may be present on the output of the receiver's FM detector. A small voltage gain is also provided by this filter, and a positive DC offset voltage from the host FM receiver is required for its proper operation.

The output of the low-pass filter is then applied to two precision peak detectors and a data slicer. The purpose of this circuitry is to convert the sinusoidal FSK waveform delivered by the receiver to a binary waveform compatible with digital logic circuitry without being affected by the DC offset voltage riding along with the FSK waveform.

A positive peak detector is formed around U1B, D1, and U2B, while a negative peak detector is formed around U1C, D2, and U1D. In operation, the positive peak detector provides an output voltage equal to the maximum voltage excursion of the filtered FSK voltage waveform, while the negative peak detector provides an output voltage equal to the minimum voltage excursion.

Potentiometer R7 allows the average of the two peak voltages ((positive peak + negative peak) / 2), which is a voltage level equal to the exact center of the filtered FSK voltage waveform, to be accurately set and used as a threshold voltage for the data slicer. The beauty of this arrangement is that if the FSK's DC offset voltage should rise or fall due to receiver mistuning or the effects of doppler shift, the average voltage prcoduced by the peak averaging circuit will rise or fall accordingly and always remain exactly at a level equal to the precise midpoint of the filtered FSK voltage waveform, thereby insuring proper operation of the data slicer. Essentially, what this circuitry does is subtract the DC offset voltage from the incoming signal (FSK+DC), and yield a clean FSK signal that can then be further processed by the modem.

The data slicer is simply a voltage comparator designed around operational amplifier U2A that provides a binary output voltage that is a function of the peak FSK input voltage. The comparator's output voltage goes high if the received FSK peak positive voltage is above the threshold voltage, or goes low if the received FSK peak negative is below the threshold voltage.

The output of the data slicer is then fed through exclusive-OR (XOR) gate U3C which acts as a buffer to produce a high amplitude 12-volt peak-to-peak voltage waveform with sharp rise times and fall times. The waveform at this point is essentially a level-converted binary representation of the received FSK transmission, and could be applied directly to a packet radio terminal node controller for decoding if it were not for the fact that the 9600 baud data received was randomized prior to transmission and must first be unscrambled before processing can take place by the host TNC.

Bit Clock Detection and Regeneration

Linear feedback shift register (LFSR) unscrambling circuitry must be driven by clock pulses that match the frequency and phase of the clock in the transmitting TNC. Since a 9600 baud clock signal is not directly available from the FSK transmitter, it must be regenerated locally from the received FSK data itself using additional circuitry in the modem.

In order to accomplish this, level converted FSK data available from U3C is applied to an edge detector consisting of an exclusive-OR gate (U6C) and an RC timing network. The edge detector produces an output pulse for every FSK logic level transition received. This waveform is known as a protoclock, and is in phase with the transmitting station's bit clock. It is, however, non-continuous, and so is used to drive a phase locked loop (PLL) to produce a continuous clock signal. The phase locked loop is designed around a micropower 4046B CMOS PLL (U8), and operates at 9600 Hz. The regenerated clock, which is available on pin 4 of U8, is used to drive the LFSR unscrambler as well as a sample and hold circuit that is part of a unique FSK data carrier detection (DCD) circuit.

The center frequency of the phase locked loop's VCO is set to 9600 Hz by R23. Resistors R24, R25, and capacitors C13 and C14 form a loop filter that allows the phase lock loop to acquire and maintain lock with the received FSK signal despite small amounts of noise that may be present with the incoming signal.

Low-Pass Filtering

The level converted FSK data available from U3C is further processed through a fifth-order low-pass filter to match the modem's detector bandwidth to that of the transmitted signal. The filter also delays the FSK waveform in time so that by the time the signal reaches the output of the filter, the bit centers of the filtered FSK waveform are synchronous with the rising edges of the regenerated clock pulses and can be sampled at the optimum moment by D-Latch U5B prior to final processing by the unscrambler. The filter is designed around transistor Q1, operational amplifiers U2C and U2D, and their associated RC passive components.

The output of the filter is available on pin 14 of U2D, and if the waveform present at this point were viewed on an oscilloscope whose horizontal sweep were triggered on the regenerated FSK clock waveform, an "eye diagram" would be observed. The "eye," or filtered bit center, is widest at the modem's sampling point (see Figure 9). The filtered waveform is converted back to a digital logic compatible square waveshape through comparator U4C, and applied to an unscrambler consisting of a 4013B D-Latch (U5), a 4006B shift register, and three exclusive-OR gates. The output of the unscrambler is converted to 0 to 5 volt TTL levels by Q2, and is made available for final processing by the host TNC.

Data Carrier Detection

Filtered FSK data is also processed through a sample-and-hold circuit consisting of operational amplifiers U9A, U9B, and a 40168 CMOS bilateral switch (U10A). The switch is triggered for a very brief instant at the center of the filtered FSK waveform by a differentiation network consisting of capacitor C15 and R26. 9600 samples of the voltages taken at the filtered waveform bit centers are taken and stored in capacitor C16 every second. The voltage present across C16 is buffered through voltage follower U9B.

The output of the sample-and-hold circuit feeds an LM3914 LED dot/bar display driver, which forms the basis of the modem's data carrier detection circuitry. The LM3914 has 10 outputs, each of which represent a discrete voltage level of the sampled FSK sampled bit center. The presence and quality of an FSK signal can be determined by examining the filtered bit center. A bit center that is wide and noise-free is an indication of reception of a valid, high quality FSK signal. Noise and other distortion narrows the gap within the opening of the bit center. A signal that is pure noise will display no opening at all.

The LM3914's four center outputs represent voltages corresponding to the central region of the filtered FSK bit centers. These outputs are logically "ORed" together and converted to a ground-referenced voltage through transistor Q3. The upper three and lower three outputs of the LM3914 are also combined together through transistor Q4 to produce a reference voltage against which the center outputs can be compared. The upper and lower outputs remain fairly constant regardless of whether noise or a clean FSK signal are being received. The difference between the two transistor output voltages is an indication of the signal-to-noise ratio of the received FSK signal. This voltage ratio is used to drive signal quality meter, M1, as well as a voltage comparator designed around operational amplifier U9C.

The voltage comparator generates an FSK Data Carrier Detect (DCD) output signal for use by the host TNC. Hysteresis and some low-pass filtering is used around the comparator to bolster its immunity against noise or false signals. The DCD output voltage logic is active low to match the requirements of most terminal node controllers. A DCD connection to the TNC is required for half-duplex communications, but required for full-duplex satellite communications. LED D4 provides the user with a visual indication of FSK data carrier detection.

Automatic Frequency Control

The slicer threshold voltage available at the center of potentiometer R7 is applied to a second voltage comparator designed around operational amplifier U4D. This comparator compares the FSK DC offset voltage with that of a regulated reference voltage adjustable through potentiometer R8. Since the offset voltage present at R7 drifts slowly with time due to doppler shift, the comparator output can be used to tune the downlink receiver lower in frequency to compensate for the effects of doppler shift.

The AFC system used in the KD2BD 9600 Baud Modem is designed to connect to transceivers that allow UP/DOWN tuning through the front panel microphone connector. The modem sends a series of pulses generated by operational amplifier U9D to the groundstation receiver to tune it lower in frequency during a satellite pass in compensation for doppler shift. Since the direction the FSK DC offset voltage changes with receiver tuning may differ between receivers of different makes and models, an AFC polarity inverter circuit consisting of CMOS switches U10B, U10C, and U10D is installed between the output of U4D and the gated pulse generator, U9D. The inverter may be bypassed through switch SW1.

A selection of positive or negative tuning polarities is provided by the modem as well. Yaesu transceivers require a 5 volt pulse on the frequency control line to tune the receiver from the microphone connector, while others require a switch to ground. Potentiometer R8 is adjusted so that the AFC circuit properly adjusts the downlink receiver when it is incorrectly tuned, and becomes inactive after proper compensation is applied by the modem and the receiver is properly tuned. Since doppler shift causes the signal received from an earth orbiting satellite to drift lower in frequency and never higher, the automatic frequency control system used in the KD2BD 9600 Baud Modem tunes the downlink receiver in one direction only.

LED D3 provides a visual indication of when the automatic frequency control circuitry in the modem is active. The LED flashes as frequency corrections are made to the downlink receiver during a satellite pass.

FSK Generation

As stated earlier, 9600 baud FSK data is randomized or "scrambled" prior to transmission. As in the case of the LFSR unscrambler used in the receive section of this modem, a clock at 9600 Hz is required for proper operation of the LFSR scrambler circuitry. However, the TX Clock available from many TNC modem disconnect headers is at 16 times the transmitted data rate (153,600 Hz). The x16 clock frequency must therefore be divided by 16 to yield a proper signal for operating the scrambler.

U14, a CMOS 4040B binary counter, is used to divide the x16 TX Clock down to 9,600 Hz. The counter is synchronized to the TX Data waveform through a differentiation network consisting of capacitor C26 and resistor R59. The TTL digital logic levels from the TNC are also converted to the 0 to +12 volt CMOS logic levels required by the modem through transistor Q7, DC blocking capacitor C25, and resistors R57 and R58.

The LFSR scrambler is similar in design and construction to the unscrambler used in the receive section of this modem. Scrambled transmit data is available on pin 4 of U3B, and is processed through a six pole low-pass filter designed around transistor Q8, operational amplifiers U4A, U4B, and their associated passive RC components. The output of the filter available on U4B pin 7 is a pure raised cosine voltage waveform. The waveform is then attenuated and level converted to match and amplitude and DC bias level required by the varactor diode in the transmitter's FM modulator to produce a peak carrier deviation of 3.5 kHz.

Since a direct connection is used between the modem the transmitter's varactor diode, the modem output voltage levels control both the FSK modulation level as well as the FSK transmitter's center frequency. Potentiometer R71 adjusts the DC bias voltage level added to the modem's FSK output waveform and sets the transmitter's center frequency. R70 sets both the center frequency and the peak FM deviation level of the FSK transmitter. These adjustments are not mutually exclusive, so some skill and patience are required to set each one properly.

Modem Construction

The KD2BD 9600 Baud Modem was constructed on a single perforated circuit board measuring 4.5" x 4.5" inches (11.5 x 11.5 cm) using point-to-point wiring between individual component leads. Perforated circuit board construction is a simple, yet effective method of prototyping electronic circuit designs, and takes only slightly greater construction time compared to building a kit from a printed circuit board design.

Integrated circuit placement is shown in Figure 10. This layout is by no means sacred, but is included as a guideline for construction. All dual in-line package (DIP) integrated circuits should be installed in low-profile sockets to ease construction and allow initial circuit testing to be performed without the risk of damaging any of the ICs used in the modem. The two low-power voltage regulator chips and all transistors may be safely wired in without the use of sockets.

TNC Interfacing

The KD2BD 9600 Baud Modem was designed to interface with an MFJ model 1270B terminal node controller (TNC). The 1270B is a TAPR TNC-2 clone, and as such, uses a standard modem disconnect header arrangement to interface with external modems. Figure 11 shows how connections are made between the KD2BD 9600 Baud Modem and the MFJ-1270B or other TNC-2 clone. Users of TNCs by other manufacturers should consult technical literature pertaining to their specific TNC to determine the proper modem disconnect header connections that need to be made between the TNC and an external modem such as the one described here.

Alignment and Testing

After all circuit components and IC sockets have been wired on the circuit board, alignment and testing of the modem can begin. Before installing any integrated circuits into their sockets, verify that connections made for +Vcc and GND are correct for each chip. Table 2 indicates the DC power connections for each integrated circuit in the modem, and may be used as a reference. This test may be made using an ohmmeter. Also verify that a short does not exist between +Vcc and GND.

If all connections check out correctly, apply 12 volts DC to the modem and verify that the output voltage of each voltage regulator is correct and that the proper +Vcc voltage is being fed to each chip socket using a voltmeter. Table 2 may again be used as a reference. U8 is the only chip that is supplied with +8 volts. All others are supplied with 12. If all is found to be correct, power can be removed, and all chips may be carefully inserted into their proper sockets.

Connect the modem to the host TNC and connect power to the modem. Adjust both R70, the FSK deviation control, and R71, the FSK Center Frequency adjust control, to mid position. An oscilloscope attached to the FSK Output connection of the modem should display a pseudo-random serial data stream. If a high impedance audio amplifier and speaker or audio signal tracer is placed at this point, a hissing sound should be heard if the modulator is working properly.

Audio loop back testing can be accomplished by connecting the modem's FSK Output to the modem's FSK Input connection. The permits alignment and testing of the demodulator using the modulator section of the modem as a test signal source. With a high impedance voltmeter, measure the DC voltage at U1A pin 1 with respect to ground. Adjust the Slicer Level control, R7, until the voltage seen at U2A pin 2 is the same as what is seen at U1A pin 1. Temporarily remove the audio loop back connection, and connect a frequency counter to pin 3 of U8. Adjust R23 until the counter reads 9600 Hz. Re-connect the audio loop back connection. At this point, the Data Carrier Detect (DCD) LED should be lit and the Signal Quality meter should show significant up scale deflection.

With a dumb terminal or computer running a terminal emulation software package connected to the TNC, set the TNC's full-duplex parameter ON (FULLDUP ON). Proper operation of the modem may be had by trying to establish a connection with the callsign assigned to your TNC. The throughput experienced should be flawless.

Transceiver Interfacing

Proper interfacing between any 9600 baud FSK modem and transceiver will vary depending on the transceiver circuit design and the make and model of the transceiver. Mike Curtis, WD6EHR, authored a "9600 Baud Packet Handbook" that describes 9600 baud packet communications, including modem connection points as well as modifications to popular transceivers. Mike's handbook was widely distributed electronically via packet radio BBSs several years ago, and is probably available today via the Internet. Many Internet sites contain 9600 baud circuit modifications and connection points for 9600 baud modems for many different types of transceivers. The proper transceiver interfacing methods or transceiver circuit modifications are left to the expertise of the reader, but the following general information should be of use regardless of the specifics of the radio transceiver used in conjunction with this modem.

FSK signals are tapped off from the receiver's FM detector prior to any de-emphasis circuits or DC blocking capacitors. The input impedance of the KD2BD 9600 Baud Modem is high, so the modem should not load down the FM detector and cause distortion or lack of sensitivity when receiving FM voice signals while remaining connected to the transceiver as is the case with some other modem designs.

Transmit data from the modem is injected into the varactor diode associated with the transmitter's modulator through an RF isolation network. Figure 12 shows a representative isolation network consisting of a 47k ohm resistor and 100 pf capacitor. The isolation network allows signals from the modem to control the capacitance of the varactor diode while preventing RF that appears across the diode from being fed back into the modem.

True frequency modulation is produced when the varactor diode used as the FM modulator is connected in series or parallel with a crystal whose associated oscillator directly affects the transmitter's operating frequency. Transmitters that modulate a varactor diode associated with a phase locked loop (PLL) voltage controlled oscillator (VCO) cannot produce FM at low modulating frequencies and should not be DC coupled to this or any other 9600 baud modem. Transmitters that produce phase modulation will not produce high-quality FSK signals, and should be avoided.

Varactor diodes used for FM modulation are typically supplied a temperature-compensated DC operating bias voltage in addition to transmit audio. The exact method used to do this varies from manufacturer to manufacturer. The DC output voltage of the modem must match the DC bias voltage present on the varactor diode so as not to alter its operating bias or affect the center frequency of the transmitted signal. Potentiometer R71 adjusts the modem's DC bias output voltage level. This level is also affected to some extent by R70, the Deviation Level Control. The proper setting of these controls can be easily made by determining the DC bias voltage present on the varactor diode and the sensitivity of the FM modulator.

Determining Varactor Sensitivity

A 10k linear taper potentiometer placed across a 9 or 12 volt DC voltage source can be used to vary the bias voltage across a varactor diode and determine how the changing bias voltage effects the final operating frequency of the transmitter. The negative end of the battery and potentiometer combination should be connected to the transceiver's ground, and the wiper of the pot can be connected to the varactor diode through an isolation network such as the one shown in Figure 12. Varying the potentiometer will directly vary the operating frequency of the transmitter, and a voltmeter placed between the wiper of the pot and ground will measure the voltage being applied to the varactor.

Using a frequency counter to measure the operating frequency of the transmitter, first determine and record the DC voltage required for the transmitter to produce an RF carrier 3.5 kHz below what the transmitter is set to, and then determine and record the DC voltage required to produce an RF carrier 3.5 kHz above what the transmitter is set.

Attach a DC-coupled oscilloscope to the output of the modem. With the modem powered and attached to the host TNC, adjust the Deviation and FSK Center Frequency controls to produce a waveform with a peak-to-peak voltage to equal to the minimum and maximum voltages determined in the varactor sensitivity tests. Once these controls are properly set, the modem may be connected to transmitter, and the result will be a properly centered FSK signal with a peak carrier deviation of +/- 3.5 kHz.

Modem Operation

If the modem is to be used for terrestrial packet radio communications, there is no need to make automatic frequency control (AFC) connections between the modem and the RF transceiver. The same is true if the modem is to be used for satellite communications and the frequency of the ground station receiver is under the control of satellite tracking and Doppler correcting software. If corrections for Doppler shift are not made, then the modem's automatic frequency control feature may be used to keep the ground station receiver properly tuned to the satellite's downlink transmissions.

Two separate digital AFC signal polarities are provided by the modem. Please consult the operating manual of the transceiver used in conjunction with this modem to determine whether a positive pulse or a switch to ground is necessary to tune the receiver lower in frequency via the microphone connection, and make the appropriate connections between the modem and the ground station transceiver.

The AFC Adjust control, R8, may be properly set by tuning to an unmodulated carrier, and adjusting the control until the voltages on pins 12 and 13 of U4D are equal, or the voltage on pin 14 just toggles between a voltage close to +Vcc and a voltage close to ground.

Proper tuning of the ground station transceiver may require manipulation of switch SW1, depending on the heterodyne scheme of the transceiver used in conjunction with the modem. Once the proper setting is determined, the switch needs no further adjustment unless the transceiver is changed to a different make or model.

Switch SW2 needs to be front panel mounted, and is used to turn the automatic frequency control feature on and off. FSK signals received from satellites should be first tuned in manually with the AFC turned off. After the signal is acquired, the AFC can be turned on and the modem will track the downlink signal for the remainder of the satellite pass. Proper tuning is indicated when the highest upscale deflection of the Signal Quality meter is achieved.

Communication with any of the current 9600 baud satellites employing the FTL0 file transfer protocol (such as UoSAT-OSCAR-22, KITSAT-OSCAR-23, KITSAT-OSCAR-25, or TMSAT-OSCAR-31) requires that the computer connected to the TNC used at the ground station run FTL0 client software such as PB/PG, The Microsat Software Ground station Software Suite (MSGSS), WiSP, or equivalent software. Pacsat satellite communication software may be found on the Internet at: Pacsat ground station software is not, however, required to communicate with the packet radio bulletin board carried on-board the FUJI-OSCAR-29 satellite or the Mir space station.

Cost Saving Measures

The cost of commercial radio communication equipment can be quite prohibitive to prospective amateur satellite operators. While modern transceivers provide many versatile functions and features, few of these features are required for digital satellite communication. Considerable money can be saved by taking a more simplistic approach to amateur satellite communications rather than solving problems with a checkbook.

Considering the fact that most 9600 baud OSCAR satellites have one or two uplink channels in the 2-meter FM band, a sophisticated uplink transmitter is not required to access the 9600 baud satellites. An old 2-meter mobile FM transceiver or a Hamtronics VHF-FM exciter may be modified to produce FSK along with this modem. Downlink reception is possible using a modified programmable UHF scanner (along with a low-noise preamplifier). Ed Krome, K9EK, described an effective and low-cost method of receiving 9600 baud satellite downlink signals using a 70-cm to 10-meter downconverter feeding a Ramsey Electronics 10-meter FM tunable receiver kit in the March/April 1992 issue of the AMSAT Journal. The bottom line is that amateur satellite communications does not need to be expensive. Some technical savvy and a little "ham ingenuity" can go a long way towards saving many thousands of hard-earned dollars.

Higher Speeds

Although not tested, it should be a simple matter of scaling the RC time constant of the PLL and increasing the cutoff frequency of the low-pass filters to permit the KD2BD 9600 Baud Modem to operate at much higher data rates. Of course higher data rates would require a receiver bandwidth in excess of that available from a narrowband FM receiver designed for voice communications. However, it isn't a difficult matter to design and build wideband FM receivers these days with the wide variety of single chip FM receivers currently available to experimenters. A simple 10-meter wideband FM receiver can easily be designed around a Motorola MC3362 chip, for example, and this can be preceded by a 70-cm downconverter to allow reception of high-speed data with little difficulty.


Digital OSCAR satellites bring worldwide communications to all corners of the globe at low cost, and even permit unattended and automated ground station operation. Many digital OSCAR satellites not only function as store-and-forward message switches, but also carry earth imaging cameras and scientific experiments that survey the near-earth environment.

The KD2BD 9600 Baud Modem brings to the Amateur Radio community a new hardware design capable of providing high performance 9600 baud packet radio communications that permits access to these exciting digital satellites for less than the cost of a desk microphone. It is hoped that the KD2BD 9600 Baud Modem will find good use in amateur radio stations around the world, and that its design will promote a further understanding of the digital communication methods used in the Amateur Radio Service, and foster increased experimentation, development, and refinement of communications techniques used in both terrestrial and extra-terrestrial communications.

See you on the birds!


John A. Magliacane, KD2BD